Switching helix power supply for a TWT

ABSTRACT

Switching power supply for generating a voltage for a pulsating load (2), in particular for generating a helix voltage for a TWT. The switching power supply is provided with a dc voltage source (1), a buffer (12) from which the load (2) is powered, and switches (3) and a control circuit (5) for regulating the charging of the buffer from the dc voltage source (1), whereby the power supply is provided with a circuit coupled to the dc voltage source (1), which circuit consists of a current source (6), the above-mentioned switches (3) and a primary (9) of a converter (7). The buffer (12) is powered from the secondary (11) of the said converter (7). The control circuit (5) controls the switches (3) by means of a signal which is a function of the rhythm of the pulsating load and the voltage across the buffer.

BACKGROUND OF THE INVENTION

The invention relates to a switching power supply for generating avoltage for a pulsating load, in particular for generating a helixvoltage for a TWT, where the switching power supply is provided with adc voltage source, a buffer from which the load is powered, and switchesand a control circuit for regulating the charging of the buffer from thedc voltage source.

The phase performance of a transmitter provided with a TWT is directlydependent on the helix voltage of the TWT. If such a transmitter is usedin radar equipment, it is of crucial importance that the phaseperformance of the transmitter is extremely accurate. After all, Dopplerinformation of a target is obtained from the phase difference betweentransmitted and reflected radio waves. This means that the supplyvoltage for the helix power of a TWT must be extremely accurate. Aswitching supply as described above however is not sufficientlyaccurate. The lack of accuracy in the power supply is caused by the factthat the buffer is charged in steps by switching of the supply. The sizeof such a step therefore contributes to the inaccuracy of the powersupply.

SUMMARY OF THE INVENTION

The present invention has for its object to provide the possibility ofdeveloping a particularly accurate helix supply by means of a switchingpower supply provided with a circuit, coupled to the dc voltage source,which circuit consists of a current source, the above-mentionedswitches, and a primary of a converter, where the buffer is powered fromthe secondary of the said converter and where the control circuitcontrols the switches by means of a signal which is a function of therhythm of the pulsating load and the voltage across the buffer.

Because switching of the supply is a function of the possibly staggeredPRF of the TWT, it is possible to charge the supply buffer in one go.This implies that, according to the invention, the supply is extremelyaccurate because the accuracy of the supply is not impaired by the stepsize as described above. In this context, accuracy means the extent ofthe TWT cathode voltage variation from pulse to pulse. The accuracy ofthe supply is now a function of the accuracy of a circuit included inthe control circuit for measuring the voltage across the buffer. Aspecial embodiment of the voltage measurement circuit is describedbelow. This embodiment can further increase the accuracy of the powersupply.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be explained with reference to the accompanyingfigures, of which:

FIG. 1 is an embodiment of the power supply according to the invention;

FIGS. 2A-2C are characteristics for explaining the operation of thepower supply according to the invention;

FIG. 3 is a first special embodiment of the power supply according tothe invention;

FIG. 4 is a second special embodiment of the power supply according tothe invention;

FIG. 5 is an embodiment of the control circuit of the power supply.

DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 1 illustrates a dc voltage source 1 which supplies the power forthe helix of a TWT 2. Switches 3A and 3B can be closed via lines 4A and4B under control of control circuit 5. When switches 3A and 3B areclosed simultaneously, a current source 6 will supply a constant currentI_(s). The current I_(s) runs from the positive terminal of dc voltagesource 1 via the said switches and via a current converter 7 to thenegative terminal of dc power supply 1. Current converter 7 consists ofa high-voltage transformer 8 of which the primary 9 is fed with currentI_(s) and of a diode 10 which is fed from secondary 11 of high-voltagetransformer 8. Current I_(s) through primary 9 determines the primaryvoltage of transformer 8 and thus current I₁ through secondary 11. Abuffer 12 is charged via diode 10 by current I₁ of secondary 11.Charging current I₁ of buffer 12 is thus directly dependent on the valueof current I₁ through primary 9.

The winding ratio N_(p) /N_(s) of the primary and secondary (9 and 11respectively) of transformer 8 is such that the voltage for chargingbuffer 12 will be sufficiently high. If switches 3A and 3B are opened bycontrol means 5, current I_(s) through the primary will become zero. Themagnetisation energy in transformer 8 can then be returned to dc voltagesource 1 via diodes 13A and 13B.

The properties of the control circuit will be discussed below withreference to FIGS. 2A, 2B and 2C. The TWT 2 is controlled via line 14 bya grid modulator which is not indicated in FIG. 1. Between time t=t₀ andtime t=t₁ TWT 2, under control of line 14, generates a pulse with apower P_(z) as indicated in FIG. 2A. For this purpose, the TWT drawsenergy from buffer 12. Voltage |V_(b) | will as a result decrease asindicated in FIG. 2B. The pulse is terminated at time t=t₁, so thebuffer is no longer discharged. At this time t=t₁, switches 3A and 3Bare also closed, causing buffer capacitor 12 is recharged with aconstant current

    I.sub.1 =I.sub.s ·N.sub.p /N.sub.s.

As a result of charging current I₁ voltage V_(b) across buffer 12 willincrease again. Voltage V_(b) is measured by control circuit 5 via lines15A and 15B. A special embodiment of control circuit 5, enabling veryaccurate measurement of the said voltage, is later described withreference to FIG. 5. Control circuit 5 makes sure that switches 3A and3B are opened again as soon as V_(b) =V_(ref1).

A reference voltage -V_(ref1) /M is supplied to control circuit 5 vialine 16, determining the rating of voltage V_(b) when buffer 12 is fullycharged. M is a predetermined constant with M>>1.

At time t=t₂, V_(b) =V_(ref1), causing switches 3A and 3B to open andcharging current I₁ to become 0, see FIGS. 2B and 2C. Between t=t₃ andt=t₄, TWT 2 is triggered via line 14 to transmit a short pulse, aso-called follow-up pulse (see FIG. 2A). Voltage V_(b) of buffer 12 willnow decrease less because the buffer is discharged during a shorterperiod of time. Control circuit 5 ensures that buffer 12 is rechargedbetween times t=t₄ and t=t₅ in the same way as described above betweent=t₁ and t=t₂.

It may sometimes be advisable to refrain from transmitting radar pulses,e.g. to prevent the radar installation in question from being located.The result is that buffer 12 is not regularly discharged andsubsequently recharged. When this happens, voltage V_(b) across buffer12 may slowly decrease as a result of a leakage current. However, assoon as the control circuit establishes that voltage V_(b) is lower thanV_(ref2), it will close switches 3A and 3B, so that the buffer will berecharged. As soon as V_(b) =V_(ref1), switches 3A and 3B will be openedagain. FIG. 2 illustrates a situation in which during a long period oftime (between t₅ and t₆) no radar pulses are transmitted, as a result ofwhich voltage V_(b) slowly decreases. In this situation t=t₆ is thepoint in time when V_(b) =V_(ref2), and t=t₇ is the point in time whenV_(b) =V_(ref1).

FIG. 3 shows an embodiment of the power supply in which the currentsource is provided with a PNP transistor 16, a resistor 17 and areference voltage source 18 for generating a reference voltage V_(ref3).By means of reference voltage V_(ref3) current I_(s) can be adjusted.

The efficiency of this circuit can be determined as follows: For theamount of energy supplied to buffer 12 applies:

    W.sub.0 =V.sub.p ·I.sub.s ·(t.sub.2 -t.sub.1),

where V_(p) is the input voltage across current converter 7. For theamount of energy supplied by the dc power supply applies:

    W.sub.in =V.sub.g ·I.sub.s ·(t.sub.2 -t.sub.1),

where V_(g) is the voltage of dc power supply 1.

With V_(p) =1.75⁻¹ V_(g) as a practical value, the efficiency η is:

    η=W.sub.0 /W.sub.in ·100%=1/1.75·100%=57%.

A special embodiment of the power supply with a particularly highefficiency is shown in FIG. 4. Current source 6 consists of a currenttransducer 19, a resistor 20, a voltage source 18 and selfinduction 21.Because the current transducer secondary (winding 23) is fed with a dccurrent I_(p) =V18/R20, the said secondary is in a saturated condition.When switches 3A and 3B are closed, a voltage occurs across primarywinding 23. For the resulting current applies:

    I.sub.s =N.sub.s /N.sub.p ·I.sub.p,

where N_(s) /N_(p) is the winding ratio of current transducer 19. Aslong as switches 3A and 3B are closed, this current will keep goinguntil the current transducer on the other side of the B-H curve of thecore material is saturated. However, switches 3A and 3B cannot be closedfor that amount of time. This is prevented by the limited time duringwhich switches 3A and 3B are closed.

A certain amount of energy W₁ =(V_(g) -V_(p))·I_(s) (t₂ -t₁) is storedin selfinduction 21 during the period t₁ to t₂. As from t=t₂, an amountof energy W₂ =V_(g) ·I_(s) ·t_(r) is returned to dc power supply 1during a period of t_(r) seconds, with the result that, using conditionW₁ =W₂, applies:

    t.sub.r =(V.sub.g -V.sub.p)/V.sub.g ·(t.sub.2 -t.sub.1).

The energy losses now consist in the losses W₁₃ in diodes 13A and 13C,W₂₂ in windings 22, W₂₀ in resistor 20 and the losses W₃ in switches 3Aen 3B. For the above applies:

    W.sub.13 =2V.sub.13 ·I.sub.s ·t.sub.r

    W.sub.22 =I.sub.s.sup.2 ·R.sub.22 (t.sub.r +(t.sub.2 -t.sub.1))

    W.sub.20 =(N.sub.s /N.sub.p ·I.sub.s).sup.2 ·R.sub.20

    W.sub.3 =2I.sub.s ·V.sub.3 ·(t.sub.2 -t.sub.1),

where V₁₃ is the threshold voltage of a diode 13A or 13C and V₃ is thevoltage across a switch 3A or 3B. Using practical values for thevariates of the above formulas, a 93% efficiency can be realised. Thisespecially high efficiency is achieved mainly as a result of the lowoutput inpedance of current source 6.

FIG. 5 shows a possible embodiment of control circuit 5. Because voltageV_(b) is in the region of 30-50 kV for a helix, an accurate attentuationof the voltage will be required before the voltage is suitable forfurther processing. For this purpose, control circuit 5 is arranged insuch a way that not -V_(ref1) is used as a reference voltage but-V_(ref1) /M. A low reference voltage -V_(ref1) /M, where M>>1, isclearly much simpler to generate than V_(ref1). For this purpose, line15B is connected to earth, while line 15A is connected to an end of acircuit 24, which consists of N identical impedances 25 connected inseries. The other end of circuit 24 is connected via a coaxial cable 26with the inverting input of a operational amplifier 27. The operationalamplifier has a negative feedback with an impedance 28. Thenon-inverting input of the operational amplifier is connected to earth.A reference voltage -V_(ref1) /M is applied via resistor 29 to theinverting input of amplifier 27. The sheath of coaxial cable 26 is alsoconnected to earth, while the core of coaxial cable 26 is connected toearth on both sides via resistors 30 and 31 respectively and capacitors32 and 33 respectively. An impedance 25 consists of a parallel circuitof a resistor 34 and an impedance Z₁, where Z₁ represents a resistor 35and capacitor 36 connected in series. Impedance 28 consists of aparallel circuit of a resistor 37 and an impedance Z₂, where Z₂represents a resistor 38 and capacitor 39 connected in series. Thecircuit is dimensioned in such a way that, if V_(b) >V_(ref1), theoutput voltage of the operational amplifier V₀ >0, while, if V_(b)<V_(ref1), voltage V₀ <0.

Let us assume that every resistor 34 has a resistance value of AΩ, andresistor 29 has a resistance value of N·_(M) Ω, while resistor 37 has aresistance value of A/B Ω. Application of the second law of Kirchoff tothe junction of the inverting input of operational amplifier 27,together with the knowledge that the input current of an operationalamplifier is practically zero, leads to:

    V.sub.0 =(V.sub.b -V.sub.ref1)·(A/B/NA)=(V.sub.b -V.sub.ref1)(NB).sup.-1

where V₀ is the output voltage of operational amplifier 27. If NB>1,this formula represents the properties of a voltage divider. The acamplification factor ΔV_(b) /ΔV₀ for the signals via line 15A is Z₂₈/Z₂₅, where Z₂₈ and Z₂₅ respectively represent impedances 28 and 25. Toachieve resonance-free transmission, resistors 35, 38 and capacitors 36,39 are attuned to each other in a commonly known way to obtain:

    R.sub.35 /R.sub.38 =Z.sub.1 /Z.sub.2.

This had the advantage that to obtain tuning in accordance with thisformula, in principal no adjustment is required with respect to thestray capacitance of coaxial cable 26.

The resistance values of resistors 30 and 31 are selected the same asthe characteristic resistor of coaxial cable 26 to obtainreflection-free termination. Capacitors 32 and 33 are included to ensurethat the dc transmission is not affected by the last-mentionedresistors.

The circuit shown in FIG. 5 is especially in susceptible to interferencebecause of the low impedance of points A-B and C-D. The noise andinterference level will be low because reference voltage V_(ref1) /M isdirectly connectable, with no need for extra attenuations and, afterconnection, amplifications.

Subsequently, output voltage V₀ is supplied via line 40 to the invertinginput of a comparator 41, which is provided with a hysteresis ΔV. Forthe hysteresis applies:

    ΔV=V.sub.ref1 -V.sub.ref2 /M

The non-inverting input of comparator 41 is connected to earth. Thehysteresis is applied to ensure that control circuit 5 closes switches3A and 3B when V_(b) <V_(ref2), as indicated in FIG. 3. The logicaloutput signal of comparator 41 is supplied to a first input of aninverting OR gate. The second input of OR gate 42 is controlled by thecontrol signal (Prf) which triggers TWT 2. This ensures that switches 3Aand 3B are closed when V_(b) <V_(ref2) or when TWT 2 starts generatingan output pulse. It is also possible that the control circuit is notsupplied with the signal triggering TWT 2 because voltage V_(b) sinks toa value below V_(ref2) some time after TWT 2 starts transmitting apulse. However, OR gate 42 is used to ensure that buffer 12 is chargedat an early stage, so that it is prepared in time for the generation ofa new transmission pulse by means of TWT 2.

Finally, the output signal of OR gate 42 is supplied to two identicalamplifiers 43A and 43B, which control switches 3A and 3B via lines 4Aand 4B respectively.

I claim:
 1. A switching power supply including a dc voltage source forgenerating a dc voltage for a pulsating load comprising:buffer means forpowering said load; converter means having a primary and a secondary,said secondary connected to the buffer means; a constant current sourceconnected to the dc voltage source; switch means connected to saidconstant current source and the primary of said converter means; andload responsive control means connected to the switch means forselectively connecting said constant current source to the primary ofthe converter means in response to said pulsating load.
 2. A switchingpower supply as claimed in claim 1, wherein the constant current sourcecomprises a current transducer having a first winding and a secondwinding, the first winding being connected in series with the dc voltagesource; and control-current-generating means connected to the secondwinding for the determination of the current through the first winding.3. A switching power supply as claimed in claim 2, comprising diodemeans connecting the converter means and the first winding of thecurrent transducer to the dc voltage supply, for ensuring that when thesaid switch means is open, the energy stored in the converter andcurrent transducer can flow back to the dc voltage source.
 4. Aswitching power supply as claimed in claim 1, wherein the constantcurrent source comprises a transistor having a base supplied with areference voltage for providing a current which is a function of thereference voltage.
 5. A switching power supply as claimed in claim 2comprising diode means connected to the converter means and to theterminals of the dc voltage source in such a way that, when the saidswitch means is open, the energy stored in the converter means flowsback to the dc voltage source.
 6. A switching power supply as claimed inany of claims 2-5 and 1, in which the control means closes the switchmeans as soon as the pulsating load starts drawing energy from thebuffer, and when the buffer voltage sinks below a first reference valueand wherein the control means opens the switch means when the voltageacross the buffer rises above a second reference value.
 7. A switchingpower supply as claimed in claims 2-5 and 1, in which the convertermeans comprises rectifiers which are connected to said secondary.